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參數資料
| 型號： | LM2619MTCX |
| 廠商： | NATIONAL SEMICONDUCTOR CORP |
| 元件分類： | 穩壓器 |
| 英文描述： | 500mA Step-Down DC-DC Converter |
| 中文描述： | 1.1 A SWITCHING REGULATOR, 1000 kHz SWITCHING FREQ-MAX, PDSO14 |
| 封裝： | MO-153AB, TSSOP-14 |
| 文件頁數： | 13/16頁 |
| 文件大小： | 799K |
| 代理商： | LM2619MTCX |
Device Information
(Continued)
SOFT-START
The LM2619MTC has soft start to reduce current inrush
during power-up and startup. This reduces stress on the
LM2619MTC and external components. It also reduces star-
tup transients on the power source. Soft start is implemented
by ramping up the reference input to the error amplifier of the
LM2619MTC to gradually increase the output voltage.
THERMAL SHUTDOWN PROTECTION
The LM2619MTC has a thermal shutdown protection func-
tion to protect itself from short-term misuse and overload
conditions. When the junction temperature exceeds 150C
the device turns off the output stage and when the tempera-
ture drops below 130C it initiates a soft start cycle. Pro-
longed operation in thermal shutdown conditions may dam-
age the device and is considered bad practice.
Application Information
SETTING THE OUTPUT VOLTAGE
The LM2619MTC can be used with external feedback resis-
tors to set the output voltage. Select the value of R2 to allow
atleast 100 times the feedback pin bias current to flow
through it.
V
OUT
= V
FB
(1+R1/R2)
EXTERNAL COMPENSATION
The LM2619MTC uses external components connected to
the EANEG and EAOUT pins to compensate the regulator
(
Figure 4
). Typically, all that is required is a series connection
of one capacitor (C4) and one resistor (R3). A capacitor (C5)
can be connected across the EANEG and EAOUT pins to
improve the noise immunity of the loop. C5 reacts with R3 to
give a high frequency pole. C4 reacts with the high open loop
gain of the error amplifier and the resistance at the EANEG
pin to create the dominant pole for the system, while R3 and
C4 react to create a zero in the frequency response. The
pole rolls off the loop gain, to give a bandwidth somewhere
between 10kHz and 50kHz, this avoids a 100kHz parasitic
pole contributed by the current mode controller. Typical val-
ues in the 220pF to 1nF (C4) range are recommended to
create a pole on the order of 10Hz or less.
The next dominant pole in the system is formed by the output
capacitance (C2) and the parallel combination of the load
resistance and the effective output resistance of the regula-
tor. This combined resistance (Ro) is dominated by the small
signal output resistance, which is typically in the range of 3
to 15
. The exact value of this resistance, and therefore this
load pole depends on the steady state duty cycle and the
internal ramp value. Ideally we want the zero formed by R3
and C4 to cancel this load pole, such that R3=RoC2/C4. Due
to the large variation in Ro, this ideal case can only be
achieved at one operating condition. Therefore a compro-
mise of about 5
for Ro should be used to determine a
starting value for R3. This value can then be optimized on
the bench to give the best transient response to load
changes, under all conditions. Typical values are 10pF for
C5, 220pF to 1nF for C4 and 22K to 100K for R3.
A
O
= 20000 , Open loop gain of error amplifier
R
f
= 1 , Transresistance of output stage
M
c
= 362000 A/s , Corrective ramp slope
D = VOUT/VIN , D’ = 1-D , duty cycle
M
= (VIN - VOUT)/L1 , slope of current through inductor
during PFET on time
R
= (R1
i
R2) + 5k
, effective resistance at inverting input
of error amp
R
o
= (F
L1) / (D’
(M
c
/M
1
)+
1
2
- D)
where R
is the effective small signal output resistance of
power stage
f
P1
= 1/(2
π
A
O
R
p
C4) , low frequency pole
f
= 1/( 2
π
(Rload
i
R
o
)
C2) , pole due to Rload,Ro and
C2
f
= R
o
/ (2
π
L1) , high frequency pole from current mode
control
f
= 1/(2
π
R3
C5) , high frequency pole due to R3 and
C5
f
Z1
= 1/(2
π
R3
C4) , zero due to R3 and C4
α
= R2/(R1+ R2)
f
X
= (
α
(R
o
i
Rload)/R
f
)/(2
π
R
p
C4)
where f
gives the approximate crossover frequency.This
equation for crossover frequency assumes that f
P2
= f
Z1
.
INDUCTOR SELECTION
Use a 10μH inductor with saturation current rating higher
than the peak current rating of the device. The inductor’s
resistance should be less than 0.3
for good efficiency.
Table 1
lists suggested inductors and suppliers.
TABLE 1. Suggested Inductors and Their Suppliers
Part Number
DO1608C-103
ELL6SH100M
ELL6RH100M
CDRH5D18-100
P0770.103T
Vendor
Coilcraft
Panasonic
Panasonic
Sumida
Pulse
Phone
FAX
847-639-6400
714-373-7366
714-373-7366
847-956-0666
858-674-8100
847-639-1469
714-373-7323
714-373-7323
847-956-0702
858-674-8262
For low-cost applications, an unshielded inductor is sug-
gested. For noise critical applications, a toroidal or shielded
inductor should be used. A good practice is to lay out the
board with footprints accommodating both types for design
flexibility. This allows substitution of a low-noise shielded
inductor, in the event that noise from low-cost unshielded
models is unacceptable.
The saturation current rating is the current level beyond
which an inductor loses its inductance. Different manufactur-
ers specify the saturation current rating differently. Some
specify saturation current point to be when inductor value
falls 30% from its original value, others specify 10%. It is
always better to look at the inductance versus current curve
and make sure the inductor value doesn’t fall below 30% at
the peak current rating of the LM2619MTC. Beyond this
rating, the inductor loses its ability to limit current through the
PWM switch to a ramp. This can cause poor efficiency,
regulation errors or stress to DC-DC converters like the
L
www.national.com
13
PrintDate=2003/08/20 PrintTime=18:54:06 801627bc ds200651_p Rev. No. 1.25
cmserv
Proof
Seq=13
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